This invention relates to an apparatus and method for the dimming control of an electronic ballast for a fluorescent lamp. In particular the invention relates to an apparatus and method for such dimming control that generates low electromagnetic interference and low switching stress.
Electronic ballasts for the high-frequency operation of fluorescent lamps have been increasingly adopted as an energy efficient solution in residential, commercial and industrial lighting applications. Electronic ballasts have a number of advantages including improved efficiency of the overall system, higher lumen output per watt and longer lifetime of the fluorescent lamps. Electronic ballasts are in effect switched mode power electronic circuits, and most modem electronic ballast designs employ series resonant converters as the power circuits for driving the lamps.
FIG. 1 shows a conventional electronic ballast design. The basic concept of this design is to use the resonant voltage across the resonant capacitor Cr to cause the lamp arc to strike at high frequency, typically from 25 kHz to 50 kHz. Because of the high frequency of the excitation voltage the lamp is essentially in a continuous on-state, which provides high-quality illumination without any unwanted flickering effect.
FIG. 2 shows a conventional implementation of a half-bridge series resonant inverter for an electronic ballast application. In this arrangement the two switches S1 and S2 are complementary switches (ie when S1 is on S2 is off, and vice versa). If the potential at point Y is taken as the zero voltage reference point, then voltage Vxy will have the values xc2x1Vdc/2 where Vdc is the DC voltage applied to the ballast circuit either by an AC-DC converter if the power source is AC or by a DCxe2x80x94DC converter if the power source is DC. The operation of this conventional circuit will now be described for the purposes of illustration.
The two capacitors C are much larger than the resonant capacitor Cr and provide a stable DC voltage nominally at Vdc/2 at the point Y. By operating the switching frequency fsw of S1 and S2 slightly higher than the resonant frequency fr of inductor Lr and capacitor Cr the resonant load becomes inductive. If the current (iLr) in the inductor Lr is continuous, S1 and S2 can be turned on under zero-voltage. This zero-voltage switching is desirable because it reduces turn-on switching loss and minimises the electromagnetic interference (EMI) from the power switches. If additional small capacitors Cs1 and Cs2 are added as shown in FIG. 2, switches S1 and S2 can also be turned off under zero-voltage as long as the inductor current (iLr) is continuous.
Series resonant converter designs such as that shown in FIG. 2 are very popular. One reason for this popularity, for example, is that a circuit of this design can be used for a multiple lamp system simply by connecting several sets of resonant tanks and lamps across points X and Y. This flexibility greatly reduces the ballast cost per lamp.
Difficulties arise with the circuit of FIG. 2, however, when it is desired to provide a method of dimming control. Most electronic ballasts employ a nominally constant converter DC voltage and in order to control the light intensity of the fluorescent lamp dimming control is provided. Two methods of providing dimming control are commonly used in this type of ballast arrangement: duty cycle control and variation of switching frequency and these will now both be described.
The first method of dimming control is by control of the duty cycle (d) of the two switches S1 and S2. The ideal duty cycle is 0.5 but in practice the maximum d should be slightly less than 0.5 so that a small deadtime when both switches are off is provided to avoid shoot-through in S1 and S2. FIG. 3 shows typical waveforms of the gating signals of S1 and S2. By controlling the turn-on and turn-off times of the two switches the voltage applied to the series resonant circuit can be controlled. This method is not without its drawbacks however, especially at low duty-cycles, ie at low applied voltage, as will be seen from the following.
A major advantage of the circuit of FIG. 2 is that the switches can be turned on and off under zero-voltage conditions which substantially reduces EMI emission and switching stress in the power switches. However as will be seen below, if the duty cycle is too small the inductor current may become discontinuous and the zero-voltage switching conditions will be lost and the switches will suffer switching stress, leading to reduced reliability and increased EMI emission. This can be seen from the following explanation of the operating modes of the power converter which are described with reference to FIG. 4 of the accompanying drawings which schematically highlight the main current paths.
FIG. 4(a) shows a first stage in which switch S1 is ON while switch S2 is OFF and the main current path is highlighted in bold. In a second stage shown in FIG. 4(b) the two switches are OFF while Cs1 is charged up to VDC and Cs2 is discharged. When Cs2 is discharged the anti-parallel diode of S2 will start to conduct. Again the main current path is highlighted in bold. FIG. 4(c) shows this third stage in which the two switches S1 and S2 are both still OFF and the anti-parallel diode is conducting clamping the voltage across S2 to almost 0V and when the switch S2 is later turned on again it is turned on under this zero-voltage condition. However, this assumes that the inductor current is continuous. If the duty cycle is too small the inductor current may decay to zero before the switch S2 is turned on again giving the condition shown in FIG. 4(d). If the inductor current falls to zero before S2 is switched on again, the voltage across S2 is not clamped to near zero and as both switches are turned off the voltage across S2 and thus Cs2 will rise. When in the next stage S2 is turned on again the energy stored in Cs2 will be dissipated in S2 causing high discharge current and high switching loss and stress in S2.
In the next stage shown in FIG. 4(e) S2 is ON while S1 is OFF and the inductor current becomes negative. As both switches once more go to OFF, shown in FIG. 4(f), the anti-parallel diode of S1 starts to conduct clamping the voltage across S1 to near zero (FIG. 4(g)). Again, as with S2, if the duty cycle is not too small S1 will be switched on again before the inductor current decays to zero and so will be switched on while still clamped to near zero voltage, with the advantages discussed above. If the duty cycle is too small, however, the inductor current will decay to zero before S1 is switched on again causing the voltage across S1 and Cs1 to rise. When S1 is finally turned on again the energy stored in Cs1 is dissipated in S1 as discussed above with regard to S2 and with the same problems. This possibility is shown in FIG. 4(h).
Thus if dimming control by variation of duty cycle is provided, soft switching is possible provided that the inductor current is continuous. However if the duty cycle is reduced too far then the inductor current may at points in the cycle decay to zero and non-zero-voltage switching takes place with its attendant disadvantages of higher EMI emission and higher switching stress.
As an alternative to dimming control by duty cycle variation, it is also known to provide dimming control by varying the switching frequency. If the switching frequency is increased, the inductor impedance is increased and thus the inductor current is reduced. This allows the output of a fluorescent lamp to be controlled by varying the switching frequency and FIG. 5 shows the power of a 4-ft 40 W fluorescent lamp plotted against switching frequency. It can be seen that the lamp power, and therefore the intensity of the emitted light, decreases with increasing switching frequency.
Dimming control by varying switching frequency has its own disadvantages however. These include the following points:
1. If the inverter bridge is not soft-switched the switching loss of the inverter will be increased leading to reduced efficiency.
2. In order to achieve dimming control at low lamp power operation, the switching frequency range has to be very wide (eg from 25 kHz to 65 kHz) and in practice the frequency range of the magnetic cores, the gate drive circuits and electronic control circuit may all act to limit the range of dimming control.
3. Soft-switching is not easy to achieve over the entire switching frequency range. In particular, at light loads soft-switching cannot be achieved and the switching stress is large. The switching transients due to hard-switching are a major source of EMI emissions.
4. The power range of the dimming control is limited if the switching frequency range is small. A typical range of dimming control is from 100% load to 25% load.
Viewed from one broad aspect the present invention provides apparatus for controlling the power output of a fluorescent lamp comprising, an electronic ballast for driving said fluorescent lamp, power supply means for providing DC power input to said electronic ballast, and means for varying the voltage of said DC power input to said electronic ballast.
In one embodiment the power supply may comprise an AC power input followed by an AC-DC converter capable of providing a (i) power factor correction and (ii) variable DC output. Such converters may comprise a diode bridge followed by one of (a) a flyback converter, (b) a Cuk converter, (c) a SEPIC converter, (d) a Shepherd-Taylor converter, and (e) a boost converter. Preferably this front end converter uses soft-switching.
Alternatively in another embodiment the power supply may comprise a DC power input followed by a DCxe2x80x94DC converter capable of providing a variable DC output. The converter may be a step-down or a step-up/step-down converter.
Preferably the electronic ballast comprises a half-bridge series resonant inverter. The ballast preferably comprises two switches soft-switched at a constant frequency slightly higher than the resonant frequency of an inductor-capacitor tank of the ballast. The switches are preferably switched at a constant duty-cycle, preferably as close as possible to 0.5 while providing a short deadtime therebetween to prevent shoot-through.
Viewed from another broad aspect the present invention provides a method for controlling the power output of a fluorescent lamp driven by means of an electronic ballast in the form of a half-bridge resonant inverter, comprising operating said ballast at a constant duty cycle and a constant frequency and providing a variable DC power input to said ballast.